Synchronous motor drive system and synchronous motor drive method

ABSTRACT

The present disclosure is constructed on the prior art inverter architecture, a pulse code width modulation (PCWM). This is an open loop motor control system without sensing its rotor position. The present disclosure employs a closed loop method to track the optimum efficiency motor operating point directly. A bench load test is conducted to gather information for an AI type control, which includes both load angle vs. voltage command charts and power factor vs. voltage command charts, with load levels as parameters for certain frequency command ranges. This way, the optimum efficiency motor operating points are generated a priori. The AI type control is mechanized to track the optimum efficiency motor operating points.

RELATED APPLICATIONS

This application is a continuation of PCT/JP2018/040202 filed Oct. 29,2018, which claims the benefit of U.S. Provisional Application No.62/577,837 filed Oct. 27, 2017, each of which is hereby incorporated byreference herein in its entirety.

TECHNICAL FIELD

The present disclosure relates to a drive system and a drive method fora synchronous motor (e.g. a permanent magnet motor).

BACKGROUND

Induction motors have structures in which rotors do not have permanentmagnets, and are simple but inefficient. There are two classes of motorloads: stationary load use such as compressors, pumps and fans, andnon-stationary load use such as servo motors. Volume-wise, inductionmotors are dominantly employed for the stationary load use.

Vector control inverters were developed for permanent magnet motors, butdo not have functions to directly track the highest efficiency.

An inverter control operating system: Pulse Code Width Modulation (PCWM)scheme described in our prior patent, Japanese Patent No. 4482644entitled “Pulse Code Width Modulation Motor Drive System” (PatentDocument 1) is one which an open loop drives a three phase permanentmagnet AC (PMAC) motor by a sinusoidal wave signal without detecting therotor position. In the following, the invention of Japanese Patent No.4482644 is referred to as “prior patent invention.”

The prior patent invention has been made to open loop control PMACmotors for applications such as fans and compressors. The load level ofthese applications is typically known in advance. Therefore, in oneembodiment of the prior patent invention, in order to simplify thesystem, the V/F function is defined as a fixed function of the motorspeed.

One feature of one embodiment of the prior patent invention is a realtime calculation capability of updating an output sine function phaseangle and the associated pulse output width of the PWM signal everyconstant carrier signal interval of 51.2 μs (=0.2×256 μs, in which 0.2μs is the basic clock interval and 256 is the number of encoding pulseswithin the PWM modulation interval.) through use of finite state machinetechnology. This feature enables on/off modulation of the powertransistor by a sinusoidal wave signal using an ultrasonic carrierfrequency of approximately 20 kHz (=1/51.2 μs) to reduce annoyingacoustic noise peculiar to digital control. This concept is realized bydrawing a fractional sine function circle which is located inside theunit sine function circle which represents the maximum output voltage,and corresponds to the intermediate output voltage. This clearly definesthe relationship between an instantaneous amplitude value of thefractional sine function and a pulse width numeric in a unit PWM pulseinterval for the motor drive signal output. In the following, the realtime calculation procedure of the PWM signal using a unit sine functiontable and a V/F function table will be described in detail for oneembodiment of the prior patent invention.

Another feature of one embodiment of the prior patent invention is acalculation capability of performing calculation of both integers anddecimals by using double precision registers, in which carry/borrow ofthe result by rounding off. This allows more precise digital speedcontrol and increased motor drive stability.

Yet another feature of one embodiment of the prior patent invention is acapability of setting the maximum output frequency andacceleration-deceleration constant from outside through a single serialcommunication line. This capability widens the applicability of a singleASIC (Application Specific IC) chip to various motor drive applications.

Still another feature of one embodiment of the prior patent invention isemployment of a center modulation PWM signal. In a general one sidemodulation PWM signal, the inter-conduction between the upper arm andthe lower arm of the power transistor occurs once per unit PWM pulseinterval, whereas it occurs twice in the center modulation PWM signal.As a result, the ripple frequency of the current waveform of digitalcontrol becomes twice for the center modulation PWM signal compared toonce for one side modulation PWM signal. Therefore, the current waveformbecomes finer, and the motor can be driven more smoothly.

A still another feature of one embodiment of the prior patent inventionis compactness of the used hardware. In particular, the ASIC can employa small outline package (SOP) to incorporate all of the unit sinefunction table, the V/F function table, a clock generator, a PCWM signalencoder, and a PCWM signal decoder in one small package.

In the following, a detailed architectural and operationalcharacteristic of a PCWM motor drive system of the prior patentinvention will be explained. This motor drive system is characterized inthat a three phase PMAC motor is open loop controlled by a sinusoidalwave signal without detecting its rotor position. The present system isa very high cost performance motor drive system which employs anultrasonic carrier frequency of approximately 20 kHz to reduce acousticnoise, and can satisfy motor operating requirements of various uses byonly a single ASIC while keeping the hardware construction minimum.

A configuration of one embodiment of the prior patent invention will beexplained by referring to FIG. 1. An external host CPU 01 is connectedto an ASIC 06 via a UART serial bus 02. A single phase AC commercialpower supply input 03 is connected to an AC to DC converter 04, and itsAC input is converted to a DC bus voltage 05 of approximately 150 VDC orto approximately 300 VDC which is dependent on the system specification.The ASIC 06 with the input from the UART serial bus 02 performs realtime arithmetic operations and outputs gate drive input signals 07 to agate drive and power transistor circuitry 08. The gate drive and powertransistor circuitry 08 then outputs three sinusoidal wave motor drivesignals 09 to drive a three phase AC motor 10. The DC bus voltage 05 isprovided to a DC to DC step-down chopper 11 and the gate drive and powertransistor circuitry 08. The DC to DC step-down chopper 11 furtherprovides 3.3 VDC as a power supply 12 to the ASIC 06 and 5 VDC and 15VDC as power supplies 13 and 14 for control of the gate drive and powertransistor circuitry 08.

FIG. 2 is a block diagram showing the inside configuration of the ASIC06 and the gate drive and power transistor circuitry 08 according to oneembodiment of the prior patent invention. The ASIC 06 receives its inputsignal via the UART serial bus 02. An acceleration-deceleration constantkad of 2 bit hexadecimal number and a frequency multiplication factorkfmf of 5 bit hexadecimal number are input to the ASIC 06 immediatelyafter the power up. After the start of the motor, a command frequency fcof 8 bit hexadecimal number is input to the ASIC 06 as an alternateinput.

A clock generator 21, which is not shown here, operates on a crystaloscillator having a baseline frequency of 10 MHz which is connected tothe outside of the ASIC 06, and provides clock pulses of differentfrequencies or phases to blocks in the ASIC 06. Both a clock CK1 28 withthe repetition period of 200 ns and a clock CK3 29 with the repetitionperiod of 51.2 μs are provided as clock signals to the PCWM signalencoder 25 and the PCWM signal decoder 27. Similarly, the clock CK4 30with the repetition period of 51.2 μs is provided as a clock signal tothe PCWM signal decoder 27. Further, a clock CK5 31 with the repetitionperiod of approximately 3.6864 ms is provided as a clock signal to thePCWM signal encoder 25.

A unit sine function table 22 used in one embodiment of the prior patentinvention comprises values of 8 bit hexadecimal numbers corresponding tosine function values of the maximum amplitude (127 sin θ) between0°-360°. However, negative numbers are represented by one's complements.When a unit sine function table position n of the unit sine functiontable 22 is input from the PCWM signal encoder 25, a unit sine functionnumeric nu 24 is sent back to the PCWM signal encoder 25. When a machinefrequency fm of 8 bit hexadecimal number is input from the PCWM signalencoder 25, a V/F function table 23 sends a machine voltage vm of 8 bithexadecimal number back to the PCWM signal encoder 25.

The PCWM signal encoder 25 is the finite state machine which isoperational on the clock CK1 28. The PCWM signal encoder 25 updates themachine frequency fm and machine voltage vm every interrupt period ofthe clock CK5 31 by inputting the acceleration-deceleration constant kadand the frequency multiplication factor kfmf at the power up and thecommand frequency fc after the start of the motor via the UART serialbus 02. Further, it updates the unit sine function numeric nu 24 everyinterrupt period of the clock CK3 29 based on the updated machinefrequency fm and machine voltage vm and the frequency multiplicationfactor kfmf to calculate the PCWM numerics d 26 and output them to thePCWM signal decoder 27.

The PCWM signal decoder 27 operates on the clock CK1 28 and comprises aD flip-flop 32, a 7-bit up-converter 34, and a toggle flip-flop 36 whichare serially connected (see FIG. 12). The PCWM signal decoder 27 inputsthe PCWM numerics d 26 output from the PCWM signal encoder 25 everyinterrupt period of the clock CK3 29, and outputs center modulation PWMsignal numerics g 07 to the gate drive and power transistor circuitry08.

The gate drive and power transistor circuitry 08 comprises U phase upperarm 15 and lower arm 16, V phase upper arm 17 and lower arm 18, and Wphase upper arm 19 and lower arm 20, each connected in tandem for eachpair of three phases. The DC bus voltage 05 is on/off modulated by thecenter modulation PWM signals g 07 output from the PCWM signal decoder27, thereby the power transistors of the upper arms and the lower armsare activated. The three sinusoidal wave motor drive signals 09 aregenerated which correspond to the U, V and W phases, and applied to thethree phase AC motor 10.

FIG. 3 is the unit sine function table 22 used in one embodiment of theprior patent invention which is referred to by FIG. 2. The PCWM signalencoder 25 refers to this table for obtaining a unit sine functionnumeric nu 24 of 8 bit hexadecimal number, by inputting the unit sinefunction table position n of 8 bit hexadecimal number.

FIG. 4 illustrates a relationship between the fractional sine functionnumeric nf and the pulse width numeric pw in the unit PWM pulseinterval. The fractional sine function numeric nf represents theinstantaneous amplitude value and the pulse width numeric pw takes 255discrete positions. The unit sine function circle represents a circlewith the maximum radius of 127 corresponding to the maximum amplitudeemployed in one embodiment of the prior patent invention. From thecenter of this circle, 720 radial line segments extend in thecircumferential direction at 0.5 degree intervals, and each radial linesegment represents 720 different phase angles of the unit sine function.On and inside this unit sine function circle, there are 255 circles withradiuses equal to or smaller than 255 and each circle represents 255different motor output voltage levels. Mathematically, therefore, thereare 720×255 cross points representing combinations of phases andvoltages of the sinusoidal wave signal which can be taken by thisdigital machine.

Output frequency of the three sinusoidal wave motor drive signals 09 isproportional to the scanning speed of the unit sine function table 22,which is equivalent to the scanning speed of the rotating circle and isdetermined by the product of the machine frequency fm and the frequencymultiplication factor kfmf. Output voltage of the three phase sinusoidalwave motor drive signals 09 is proportional to the radius of thefractional sine function which is on or inside the unit sine functioncircle. When the three phase AC motor 10 starts to rotate after thestart of the motor, it originates with the rotating circle near thecenter of the fractional sine function circle group, and the scanningspeed of the rotating circle at this time is slow in proportion to theminimum rotation speed of the motor. Then this rotating circle graduallymoves to the outer trajectory with an accelerating speed. Finally, whenthe three phase AC motor 10 gets to the maximum speed, the rotatingcircle reaches the outermost unit sine function circle corresponding tothe maximum voltage, and rotates at the scanning speed corresponding tothe maximum rotational speed of the motor.

FIG. 5 is a graph showing a relationship between the pulse width numericpw in the unit PWM pulse interval of 51.2 μs and the instantaneousamplitude value of the fractional sine function numeric nf according toone embodiment of the prior patent invention. This relationship showsthat only integers are employed in digital machine calculations. By thisexact numerical relation, the instantaneous amplitude of fractional sinefunction numeric nf can be translated into the pulse width numeric pw inreal time every CK3 29 interrupt clock period of 51.2 μs.

FIG. 6 is a flow chart of a subprogram showing the one embodiment of theprior PCWM signal encoder 25. The machine frequency fm and the machinevoltage vm are updated every CK5 31 interrupt period (approximately3.6864 ms) in S62, S66, and S68 in the PCWM signal encoder 25. Theacceleration-deceleration constant kad and the frequency multiplicationfactor kfmf in S61 are input to the PCWM signal encoder 25 via the UARTserial bus 02 at the power up and stored in a frequency incrementregister pair D R63 and E R64 where D R63 stores null and E R64 storesan actual constant, and in a frequency multiplication factor register FR65, respectively.

The acceleration-deceleration constant kad can be selected outside themotor drive system of one embodiment of the prior patent invention inorder to conform to the acceleration-deceleration specifications ofmotors for various applications. There are four ramp speed options forthe acceleration-deceleration. The frequency multiplication factor kfmfcan also be selected externally in order to conform to the maximum driveoutput frequency specifications of motors for various applications.There are 31 options to select for the gate drive and power transistorcircuitry 08.

S62 shows that this subprogram is entered every CK5 31 interrupt periodto check if the command frequency fc which is input via the UART serialbus 02 after the power up is equal to the machine frequency fm in S63.If the command frequency fc is equal to the machine frequency fm, thesubprogram goes to the exit. Otherwise, an addition and subtractionroutine in S64 is entered where machine frequency register pair B R61and C R62 are used along with the frequency increment register pair DR63 and E R64 for the machine frequency fm update. Both the frequencyincrement register pair D R63 and E R64 and the machine frequencyregister pair B R61 and C R62 are double precision register pairs whichcan hold the decimal point numbers until the next clock CK5 31 interruptperiod for an enhancement of frequency control accuracy.

When the command frequency fc is larger than the machine frequency fm,the content of frequency increment register E R64 is added to thecontent of machine frequency register C R62 first, followed by anaddition of the content of frequency increment register D R63 and thecarry value to the content of machine frequency register B R61. When thecommand frequency fc is smaller than the machine frequency fm, thecontent of frequency increment register E R64 is subtracted from thecontent of machine frequency register C R62 first, followed by asubtraction of the content of frequency increment register D R63 and theborrow value from the content of machine frequency register B R61.

In S65, the subprogram goes to a unit sine function table 22 scanningspeed updating subprogram (FIG. 7). The content of machine frequencyregister B R61 is stored as an updated machine frequency fm in S66. InS67, the V/F function table 23 is entered using the updated machinefrequency fm where an updated machine voltage vm is obtained and storedin a machine voltage holding register G R66 in S68 and the subprogramgoes to exit.

FIG. 7 is a unit scanning speed updating subprogram of the unit sinefunction table 22 of S65 in FIG. 6. In this subprogram, a content of asine function table position increment register pair P R74 and Q R75 isupdated every clock CK5 31 interrupt period (approximately 3.6864 ms) asshown in S62 and S75 in the PCWM signal encoder 25. The content of themachine frequency register pair B R61 and C R62 is loaded in a machinefrequency holding register pair AH R71 and AL R72 in S71, and thecontent of the frequency multiplication factor register F R65, whichcontains kfmf, is loaded in a frequency multiplication factor holdingregister X R73 in S72. Then, in S73, a multiplication of A×X is carriedout. The upper 8 bits provides the integer portion and the lower 8 bitsprovides the decimal portion of the sine function table positionincrement number per 2 interrupt intervals (102.4 μs).

When the above calculation result is divided by 2 in S74, this providesa sine table position with the upper 8 bits for the integer portion andthe lower 8 bits for the decimal portion per 1 interrupt interval (51.2μs) corresponding to the unit PWM pulse interval. For example, if thecontent of the machine frequency register pair B R61 and C R62 is hff00(=255: integer portion only) and the content of the frequencymultiplication factor register F R65 is h09 (=9), the multiplicationresult is 255×9/256/2=4.4824. This is the sine function table positionincrement number per CK3 29 interrupt period corresponding to the threesinusoidal wave motor drive signals 09 output frequency of 121.6 Hz(=4.4824×100000/51.2/720), where 4.4824×1000000/51.2 is the tableposition increment number per 1 second, and 720 is the unit sine tablelength. In S75, the result obtained in S74 is loaded in the sinefunction table position increment register pair P R74 and Q R75, andthis subprogram goes to the exit.

FIG. 8 is a flow chart of a subprogram in which the unit sine functiontable position n is updated every CK3 29 interrupt period (51.2 μs) asshown in S81 and S82 in the PCWM signal encoder 25. An addition routinein S82 is entered where a sine function table position registers M R81,N R82, and L R83 are updated with the content of the sine function tableposition increment registers P R74 and Q R75. The sine function tableposition resisters M R81, N R82, and L R83 and the sine function tableposition increment register pair P R74 and Q R75 are double precisionregisters which can hold the decimal point numbers until the next CK3 29interrupt period for an enhancement of frequency control accuracy.

In S82, the content of the sine function table position incrementregister Q R75 is first added to the content of the sine function tableposition register L R83 first, followed by an addition of a content ofthe sine function table position increment register P R74 and the Carryto the content of the sine function table position register pair M R81and N R82. As a result, as shown in S83, the content of the sinefunction table position register pair M R81 and N R82 holds an updatedunit sine function table position number n in S84, a new unit sinefunction numeric nu is fetched and stored in a unit sine functionregister H.

In S85, the PCWM signal encoding subprogram shown in FIG. 9 is called.In S86, the unit sine function table position n is incremented by 480 toadvance the unit sine function table position n by 240°. The unit sinefunction table position n obtained in S86 is compared with 720 in S87,and if it does not exceed 720, the subprogram goes to S89. Otherwise,the subprogram proceeds to S88 where 720 is subtracted from the unitsine function table position n just calculated in S86 to reset the unitsine function table position n. In S82, S86 and S88 the decimal numbersare held in registers Q R75 and L R83 until the next CK3 29 interruptperiod for an enhancement of frequency control accuracy.

After S88, the subprogram goes to S89 where it is checked whether thethree phase sinusoidal motor drive signals 09 generation is completed.If it is not completed, the subprogram goes back to S84 and repeats thesame process. If it is completed, this subprogram goes to exit.

FIG. 9 is a PCWM signal encoding subprogram in which the PCWM numerics d26 are generated every CK3 29 interrupt period (51.2 μs) as shown in S81and S100 inside the PCWM signal encoder 25. In S90 the unit functionregister H R80 (see FIG. 8) is loaded in the unit sine function holdingregister A R91 as a multiplicand. Also in S91 the machine voltage vmobtained in S68 and held in the machine voltage holding register G R66is loaded in the machine voltage register X R92 which becomes amultiplier.

The polarity of the unit sine function numeric nu is determined bychecking the most significant bit of nu in S92. If it is zero or nu ispositive, S93 is entered and the multiplication of A×X is performed. Theupper 8 bits of the result are stored in a register A R93 and aremodulated on period numeric for nu>0. The lower 8 bits are stored in aregister X R94, which represents the decimal number of themultiplication, and is not used. In S94, h80 is added to the content ofthe register A R93, which becomes an overall on period numeric. In S95,the 1's complement of the register A R93 in S94 is taken to get anoverall off period numeric and the subprogram proceeds to S99.

If the most significant bit of the unit sine function numeric nu is 1 ornu is negative or zero, S96 is entered where the 1's complement of thecontent of the unit sine function holding register A R91 in S90 is takento get an off period numeric for the unit sine function numeric nu. S97performs the multiplication of A×X. The upper 8 bits of the result arestored in a register A R93 and are modulated off period numeric fornu≤0. The lower 8 bits of the result are stored in a register X R94, andis not used. In S98, h80 is added to the content of the register A R93,which becomes an overall off period numeric and the subprogram proceedsto S99.

In S99, the content of the register A R93 is a PWM one side modulationoff period numeric 2d for the upper arm 15, V phase upper arm 17, and Wphase upper arm 19, respectively. If nu>0, the output is 0≤2d<127 and ifnu≤0, 127≤2d≤255 will result. In S100, the content of the register A R93in S99 is divided by 2 to get the PCWM numerics d 26, which comprisesthree PWM center modulation off period numerics for the U-ph upper 15,V-ph upper 17, and W-ph upper 19 arms, respectively. The PCWM numerics d26 are input to the PCWM signal decoder 27.

FIG. 10 is a time chart illustrating how the PCWM numerics d 26generated in the PCWM signal encoder 25 in FIG. 9 are converted into apulse width numeric in the unit PWM pulse interval in the PCWM signaldecoder 27 according to one embodiment of the prior patent invention.The illustration shows both cases where the unit sine function numericnu is positive and negative or zero.

Let us examine now the nu>0 case using a real numeric by referring toFIG. 10. Assume nu=h3e (62) and vm=h80 (128), where decimal numbervalues are shown inside the parentheses for ease of calculation. In S93,A×X=62×128=7936. Taking the upper 8 bits of this multiplication resultleads to 7936/256=h1f (31). In S94, adding h80 to the upper 8 bits leadsto h9f (159). Then in S95, taking 1's complement of this result leads toh60 (96). In S99 of FIG. 9, 2d=96<127 is obtained. The output in S100 isthus d=h30 (48), which becomes the PWM center modulation off periodnumeric d for the upper arms 15, 17, and 19 in the gate drive and powertransistor circuitry 08.

Let us now examine the nu≤0 case assuming nu=hc1 (193) and vm=h80 (128).In S96, taking 1's complement of hc1 (193) results in h3e (62). Thus, inS97, A×X=62×128=7936. Taking the upper 8 bits of this multiplicationresult leads to 7936/256=h1f (31). In S98, adding h80 to the upper 8bits leads to h9f (159). In S99 of FIG. 9, 2d=159>127 is obtained. Theoutput in S100 is thus d=h4f (79) which becomes the PWM centermodulation off period numeric d for the upper arms 15, 17, and 19 in thegate drive and power transistor circuitry 08.

FIG. 11 is a comparison chart for the PCWM signal decode 27 outputwaveforms illustrating how the side modulation and the center modulationupper arms' signals look in the unit PWM pulse interval of 51.2 μsaccording to one embodiment of the prior patent invention. Thecomparison is made for the unit sine function numeric nu positive, zero,and negative cases.

FIG. 12 is a block diagram showing the detailed inside of the PCWMsignal decoder 27 in the ASIC 06 according to one embodiment of theprior patent invention. The PCWM signal decoder 27 updates the centermodulation PWM signals g 07 every CK3 29 interrupt period (51.2 μs). Thecenter modulation PWM signals g 07 comprise three upper arms' outputsignals 37 to drive the upper arms 15, 17, and 19 and three lower arms'output signals 38 to drive the lower arms 16, 18, and 20 which are bothlocated in the gate drive and power transistor circuitry 08.

During the past CK3 29 interrupt period, the PCWM signal decoder 27receives the PCWM numerics d 26 from the PCWM signal encoder 25 andwrites the data into a D Flip-Flop 32. As described in S100 of FIG. 9,the PCWM numerics d 26 represents the PWM center modulation front-halfoff period numeric for the upper arms 15, 17, and 19 in the gate driveand power transistor circuitry 08 (see FIG. 13 described later). Thelower arms 16, 18, and 20 are provided their signals from the oppositeoutput terminals of a Toggle F/F 36. A 7-Bit Up-Counter 34 inputs d 33at CK3 29 and d 33 at CK4 30 clock signals, and outputs a signal cry 35at the CK4 30 when the 7-Bit Up-Counter 34 reaches the full count ofh7f.

The CK1 28 is used as the clock signal for the 7-Bit Up-Counter 34 andthe Toggle F/F 36. For the 7-Bit Up-Counter 34 the numeric d 33 isloaded for the front-half off period d 26 generation and the numeric d33 for the rear-half off period d26 generation in the unit PWM pulseinterval as depicted in FIG. 13. The 7-Bit Up-Counter 34 loads 1'scomplementary numbers to generate the desired output.

The Toggle F/F 36 reverses the output polarity every time when itreceives the cry 35 from the 7-Bit Up-Counter 34 and the clocks CK3 29or CK4 30 from the clock generator 21, and generates the three upperarms' drive signals 37 and the three lower arms' drive signals 38. Thepolarities of the three lower arms' drive signals 38 are opposite to thethree upper arms' drive signals 37. For the brevity of explanation, deadtimes between the upper arms' and the lower arms' signals 37 and 38 areomitted here. The center modulation PWM signals g 07 are thus generatedand input to the gate drive and power transistor circuitry 08.

FIG. 13 shows an example of a unit PWM pulse polarity characteristic forthe upper arms' signals when the unit sine function numeric nu ispositive as a function of the normalized time in the unit PWM signalinterval according to one embodiment of the prior patent invention. Asshown in the illustration, when the unit sine function numeric nu iszero, there is no modulation and the pulse duty is 50% and d=0.25. Whenthe unit sine function numeric nu is positive, the pulse duty is morethan 50% and d<0.25. When the unit sine function numeric nu is negative,the pulse duty is less than 50% and d>0.25. Note that d+d=1. Themodulated signal portions are split into the front-half and therear-half portions.

FIG. 14 is a comparison chart showing that the inter-conduction betweenthe upper and lower arms of the power transistor creates twice per eachpower transistor interval compared to once for the side modulation PWMsignal according to one embodiment of the prior patent invention. Theexample chart shows the case where the U phase upper arm 15 and the Vphase lower arm 18 are conducting.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing an embodiment of a motor drive systemaccording to one embodiment of the prior patent invention.

FIG. 2 is a block diagram showing the detailed inside embodiment of anASIC and a gate drive and power transistor circuitry according to oneembodiment of the prior patent invention.

FIG. 3 is a unit sine function table referred to by the PCWM signalencoder when obtaining a unit sine function numeric by inputting a unitsine function table position number according to one embodiment of theprior patent invention.

FIG. 4 is an illustration showing a relation between the instantaneousamplitude value of the fractional sine function and the pulse widthnumeric in the unit PWM pulse interval according to one embodiment ofthe prior patent invention.

FIG. 5 is a functional relation graph between the pulse width numeric inthe unit PWM pulse interval and the fractional sine function numericaccording to one embodiment of the prior patent invention.

FIG. 6 is a flow chart of a subprogram showing how a machine frequencyand a machine voltage are updated every CK5 interrupt period(approximately 3.6864 ms) according to one embodiment of the priorpatent invention.

FIG. 7 is a unit sine function table scanning speed updating subprogramworking with the subprogram in FIG. 6 according to one embodiment of theprior patent invention.

FIG. 8 is a flow chart of a subprogram showing how a unit sine functiontable position number is updated every CK3 interrupt period (51.2 μs)according to one embodiment of the prior patent invention.

FIG. 9 is a PCWM signal encoding subprogram working with the subprogramin FIG. 8 in which a center modulation front-half off period numeric isupdated according to one embodiment of the prior patent invention.

FIG. 10 is a time chart illustrating how PCWM numerics generated in thePCWM signal encoder are converted into a pulse width numeric in the unitPWM pulse interval in the PCWM signal encoder according to oneembodiment of the prior patent invention.

FIG. 11 is a comparison chart for the PCWM signal decoder outputwaveforms illustrating how the side modulation and the center modulationupper arm signals look in the unit PWM pulse interval according to oneembodiment of the prior patent invention.

FIG. 12 is a block diagram showing the detailed inside of the PCWMsignal decoder in the ASIC according to one embodiment of the priorpatent invention.

FIG. 13 illustrates an example a unit PWM pulse polarity characteristicfor the upper a arms' signal when the unit sine function is positive asa function of the normalized time in the unit PWM signal intervalaccording to one embodiment of the prior patent invention.

FIG. 14 is a comparison chart showing that center modulation PWM signalcan create the power transistors' inter-conduction twice per each unitPWM pulse interval by one on/off operation of each power transistor inthe compared to once for a side modulation PWM signal according to oneembodiment of the prior patent invention.

FIG. 15 is a block diagram showing a motor drive feedback control systemusing a load angle as the primary controlled variable according to oneembodiment of the present disclosure.

FIG. 16 is a comparison chart between vector control and torque controlaccording to one embodiment of the present disclosure.

FIG. 17 is a chart of target load angle vs. voltage command with loadlevels as the parameter for certain frequency ranges obtained by a loadtest according to one embodiment of the present disclosure.

FIG. 18 is an external rotor type permanent magnet motor simplifieddrawing with the extended view according to one embodiment of thepresent disclosure.

FIG. 19 is a vector diagram showing relations among motor variablesaccording to one embodiment of the present disclosure.

FIG. 20 is a time chart of showing relations among terminal voltage,armature magnetic flux, and permanent magnetic flux, for a U-phase upperarm as the example according to one embodiment of the presentdisclosure.

FIG. 21 is a vector diagram defining a load angle as a function of motorvariables according to one embodiment of the present disclosure.

FIG. 22 is a vector diagram showing the motor torque is defined by theproduct of armature magnetic flux, permanent magnetic flux, and loadangle for a small load angle according to one embodiment of the presentdisclosure.

FIG. 23 is a control diagram and the associated time chart showing howthe control is mechanized using the load angle as a controlled variableaccording to one embodiment of the present disclosure.

FIG. 24 is a detailed block diagram showing how a control system ismechanized using a load angle as the primary controlled variableaccording to one embodiment of the present disclosure.

FIG. 25 is a chart showing relation of base voltage vs. frequencycommand as shown in FIG. 24 according to one embodiment of the presentdisclosure.

FIG. 26 is a block diagram showing an alternate motor drive feedbackcontrol system using a power factor angle as the primary controlledvariable according to another embodiment of the present disclosure.

FIG. 27 is a chart of power factor angle vs. voltage command with loadlevels as the parameter for certain frequency ranges obtained by a loadtest according to another embodiment of the present disclosure.

FIG. 28 is a control diagram when a voltage phase is advanced from acurrent phase and the associated time chart showing how the control ismechanized using the power factor as a controlled variable according toanother embodiment of the present disclosure.

FIG. 29 is a control diagram when a voltage phase is delayed from acurrent phase and the associated time chart showing how the control ismechanized using the power factor as a controlled variable according toanother embodiment of the present disclosure.

FIG. 30 is a detailed block diagram showing how a control system ismechanized using a power factor angle as the primary controlled variableaccording to another embodiment of the present disclosure.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS Example Problem to beSolved

Globally, motors consume nearly 60% of the whole electric power usage.In order to alleviate the global warming effect, realization of lowcarbon society is preached. There is enough room to reduce CO2 byenhancing the efficiencies of motors.

An object of the present disclosure is to present a synchronous motordrive system and a synchronous motor drive method with high efficiency.

Example Means for Solving the Problem

According to the first aspect of the present disclosure, a synchronousmotor drive system comprises: a synchronous motor; a load angle sensorfor measuring a load angle of the synchronous motor; and a controllerfor generating a drive signal based on an input frequency command andthe measured load angle, and supplying the drive signal to thesynchronous motor.

Here, the synchronous motor may be a permanent magnet motor, thepermanent magnet motor may comprise: a rotor comprising a permanentmagnet; and a stator comprising an armature, the synchronous motor drivesystem may further comprise a permanent magnet magnetic flux sensor fordetecting a permanent magnet magnetic flux, and the load angle sensormay measure a phase difference between an armature magnetic flux and thepermanent magnet magnetic flux to measure the load angle.

Here, the controller may send an armature magnetic flux phase signalrepresenting a phase of the armature magnetic flux to the load anglesensor, the permanent magnet magnetic flux sensor may send a permanentmagnet magnetic flux phase signal representing a phase of the permanentmagnet magnetic flux to the load angle sensor, and the load angle sensormay measure the phase difference between the armature magnetic flux andthe permanent magnet magnetic flux based on the armature magnetic fluxphase signal and the permanent magnet magnetic flux phase signal.

Here, the controller may send an on/off signal representing a magnitudeof the armature magnetic flux, as the armature magnetic flux phasesignal, and the permanent magnet magnetic flux sensor may send an on/offsignal representing a magnitude of the permanent magnet magnetic flux,as the permanent magnet magnetic flux phase signal.

Here, the controller may apply a sine wave voltage to the permanentmagnet motor, express a phase of the voltage in n ways (n is an integerequal to or greater than 2), and send n pulses to the load angle sensorduring one period of the voltage, and the load angle sensor may measurethe load angle by measuring the number of pulses which corresponds tothe phase difference between the armature magnetic flux and thepermanent magnet magnetic flux corresponds to.

Here, the permanent magnet magnetic flux sensor may be a Hall sensor.

Here, the controller may comprise: a load angle control block forgenerating a voltage command based on the frequency command and themeasured load angle to control the load angle; a PWM signal generatorfor generating a PWM signal based on the frequency command and thevoltage command; and an inverter for generating the drive signal basedon the PWM signal.

Here, the load angle control block may comprise: a voltage commandgenerator for generating the voltage command; a target load angle tablestoring a target load angle to be targeted for a frequency and a voltageapplied to the synchronous motor; a target load angle determinationblock for determining the target load angle based on the frequencycommand and the voltage command by referring to the target load angletable; and a load angle error calculator for calculating a load angleerror between the target load angle and the measured load angle, and thevoltage command generator may adjust the voltage command to be generatedbased on the load angle error.

According to the second aspect of the present disclosure, a synchronousmotor drive method for driving a synchronous motor, comprises: a step ofreceiving input of a frequency command; a load angle measurement step ofmeasuring a load angle of the synchronous motor; and a step ofgenerating a drive signal based on the frequency command and themeasured load angle, and supplying the drive signal to the synchronousmotor.

Here, the synchronous motor may be a permanent magnet motor, thepermanent magnet motor may comprise: a rotor comprising a permanentmagnet; and a stator comprising an armature, the synchronous motor drivemethod may further comprise a step of detecting a permanent magnetmagnetic flux, and the load angle measurement step may measure a phasedifference between an armature magnetic flux and the permanent magnetmagnetic flux to measure the load angle.

Here, the load angle measurement step may define an armature magneticflux axis at a position which is delayed from an armature voltage axisby π/2 in electric angle, and measure a phase difference between thearmature magnetic flux and the permanent magnet magnetic flux relativeto the armature magnetic flux axis.

According to the third aspect of the present disclosure, a synchronousmotor drive system comprises: a synchronous motor; a power factor anglesensor for measuring a power factor angle of the synchronous motor; anda controller for generating a drive signal based on an input frequencycommand and the measured power factor angle, and supplying the drivesignal to the synchronous motor.

Here, the synchronous motor drive system may further comprise a terminalcurrent detection sensor for detecting a terminal current of thesynchronous motor, and the power factor angle sensor may measure a phasedifference between a terminal voltage of the synchronous motor and theterminal current to measure the power factor angle.

Here, the controller may send a voltage phase signal representing aphase of the terminal voltage to the power factor angle sensor, theterminal current detection sensor may send a current phase signalrepresenting a phase of the terminal current to the power factor anglesensor, and the power factor angle sensor may measure the phasedifference between the terminal voltage and the terminal current basedon the voltage phase signal and the current phase signal.

Here, the controller may send an on/off signal representing a magnitudeof the terminal voltage, as the voltage phase signal, and the terminalcurrent detection sensor may send an on/off signal representing amagnitude of the terminal current, as the current phase signal.

Here, the controller may apply a sine wave voltage to the synchronousmotor, express a phase of the voltage in n ways (n is an integer equalto or greater than 2), and send n pulses to the power factor anglesensor during one period of the voltage, and the power factor anglesensor may measure the power factor angle by measuring the number ofpulses which corresponds to the phase difference between the terminalvoltage and the terminal current corresponds to.

Here, the controller may comprise: a power factor angle control blockfor generating a voltage command based on the frequency command and themeasured power factor angle to control the power factor angle; a PWMsignal generator for generating a PWM signal based on the frequencycommand and the voltage command; and an inverter for generating thedrive signal based on the PWM signal.

Here, the power factor angle control block may comprise: a voltagecommand generator for generating the voltage command; a target powerfactor angle table storing a target power factor angle to be targetedfor a frequency and a voltage applied to the synchronous motor; a targetpower factor angle determination block for determining the target powerfactor angle based on the frequency command and the voltage command byreferring to the target power factor angle table; and a power factorangle error calculator for calculating a power factor angle errorbetween the target power factor angle and the measured power factorangle, and the voltage command generator may adjust the voltage commandto be generated based on the power factor angle error.

According to the fourth aspect of the present disclosure, a synchronousmotor drive method for driving a synchronous motor, comprises: a step ofreceiving input of a frequency command; a power factor angle measurementstep of measuring a power factor angle of the synchronous motor; and astep of generating a drive signal based on the frequency command and themeasured power factor angle, and supplying the drive signal to thesynchronous motor.

Here, the synchronous motor drive method may further comprise a step ofdetecting a terminal current of the synchronous motor, and the powerfactor angle measurement step may measure a phase difference between aterminal voltage of the synchronous motor and the terminal current tomeasure the power factor angle.

Example Effect of Certain Embodiments

According to the present disclosure, a synchronous motor drive systemand a synchronous motor drive method with high efficiency can bepresented.

In the following, by referring to the figures, embodiments of thepresent disclosure will be described in detail.

First Embodiment

In the first embodiment of the present disclosure, a permanent magnetmotor (more specifically, a three phase permanent magnet motor) is usedas a synchronous motor.

The first embodiment of the present disclosure relates to an inexpensiveinverter system which can yield equal or higher efficiency compared to avector control inverter in a simpler method for a three phase permanentmagnet motor, which is characterized as a higher efficiency motor.

The present embodiment presents an inverter system for a permanentmagnet motor which directly tracks the optimal efficiency by torquecontrol using a load angle, despite its inexpensive structure.

Problems to be resolved by the present embodiment are as follows.

(1) Defining an attracting action between an electromagnet (armature) ofa motor stator and a permanent magnet of a motor rotor by using anumerical expression model of electromagnetic induction theory.

(2) Realizing a control system model in a static coordinate system whichcan drive a permanent magnet motor by sinusoidal wave to obtain theoptimal efficiency, contrary to a vector control counterpart.

(3) Defining a motor load angle as an index of efficiency control.

(4) Forming a table consisting of a set of values of load angles atwhich the motor efficiency is optimal for the number of rotations of thedrive motor in accordance with the magnitude of the load, by conductinga load test on the motor in advance, and adjusting the applied voltageon the motor to make the counted load angle value coincide with theideal value stored in the table.

(5) Conducting measurement of the motor load angle by converting thesine wave of each of the armature magnetic flux and the permanent magnetmagnetic flux to an on/off signal (e.g. signal which becomes on when thevalue is zero or more, and off when the value is less than zero), anddetecting the phase difference between the signals after the conversion.

(6) Enabling to count digitally what percentage of the wavelength at thecurrent drive frequency the magnitude of the load angle becomes.

(7) Making the control circuit simpler and smaller.

The PCWM inverter, on which the present embodiment is based, is onewhich drives the PMAC motor by sinusoidal wave signal in an open loopmode without sensing the rotor position.

The present embodiment enables the inverter system to operate constantlyat the optimal efficiency point in closed loop control by installingonly one sensor on the motor stator to detect the phase of the rotor.

The first feature of the present embodiment is to investigate anattracting action between the electromagnet (armature) of the motorstator excited by the inverter and the permanent magnet installed on therotor under the condition of the fixed rotation axis. Specifically, itis to figure out what rotating motion the rotor does by using anumerical expression model based on the electromagnetic inductiontheory, when the magnetic flux by the rotating magnetic field of theelectromagnet (armature magnetic flux) is defined as the driving sideand the magnet flux by the permanent magnet (permanent magnet magneticflux) of the rotor is defined as the tracking side.

The second feature of the present embodiment is (a) to define thearmature magnetic flux axis which is delayed from the armature voltageaxis by ¼ wavelength as the base armature magnetic flux axis, and (b) toconduct control using a fixed coordinate system (stationary coordinatesystem). The present embodiment employs a fixed coordinate system(stationary coordinate system) contrary to vector control which uses arotating coordinate system. In the PCWM method of the presentembodiment, the number of data in the sine wave 360° function table isstored as digital information of a multiple of 6. Such method isimpossible for an existing conventional inverter system.

The third feature of the present embodiment is (a) to define a loadangle as an index for realizing the optimal efficiency, and obtain thevalue of the load angle at which the efficiency is optimal in accordancewith the magnitude of the load by conducting a load test on the inverterand motor in advance, (b) to obtain the relationship between the commandvoltage and the target load angle which realizes the optimal efficiencyin the form of a table in advance, and (c) to build a model followeradaptive control system which adjusts the command voltage so that thevalue of the counted load angle coincides with the ideal value stored inthe table in a real machine.

The fourth feature of the present embodiment is to conduct measurementof the motor load angle by converting the sinusoidal wave of each of thearmature magnetic flux and the permanent magnet magnetic flux to a 50%duty on/off signal, and detecting the phase difference between thesignals after the conversion.

The fifth feature of the present embodiment is that in contrast with thefirst carrier used in the PCWM method which has a constant frequency,the period of the sine wave of the voltage applied to the motor changesin accordance with the drive frequency, and therefore by using thesecond carrier which is synchronized to the sine wave frequency, thepresent embodiment makes it possible to count digitally what percentageof the wavelength at the current drive frequency the magnitude of theload angle becomes. Especially, using the second carrier for measuringthe load angle is impossible for an existing conventional invertersystem. In the PCWM method of the present embodiment, carrier frequencycontrol is operated as digital information.

The sixth feature of the present embodiment is characterized as afull-digital construction by not using A/D converters and many sensorcircuits and consisting of fewer and less expensive parts in order tomake the control circuit simpler and smaller.

FIG. 15 illustrates a permanent magnet motor drive system (synchronousmotor drive system) in the first embodiment of the present disclosure,and represents a circuit block diagram for an efficiency control ofeither an air conditioner or a fan. The permanent magnet motor drivesystem 60 of the present embodiment comprises a controller 62, apermanent magnet motor 64, and a load angle sensor 66. In the presentembodiment, as a load 70, for example, an air conditioner or a fan isassumed. When a user inputs a room temperature command to a host CPU 50,the host CPU 50 inputs a frequency command which depends on the roomtemperature command, to the controller 62 inside the permanent magnetmotor drive system 60. The controller 62 generates a drive signal whichdepends on the frequency command, and supplies it to the permanentmagnet motor 64. The permanent magnet motor 64 operates in accordancewith the drive signal, and supplies a speed and a torque to the load 70.The load (air conditioner or fan) 70 operates in accordance with thespeed and the torque, and the room temperature changes.

In the present embodiment, a load angle is controlled by regulating avoltage in an inner loop. That is, the load angle sensor 66 measures theload angle of the permanent magnet motor 64, and supplies it to thecontroller 62. The controller 62 generates the drive signal based on thefrequency command and the measured load angle. Here, for the number ofrotations of the permanent magnet motor 64, in accordance with themagnitude of the load, there exists the value of the load angle (targetload angle) at which the efficiency of the permanent magnet motor 64becomes optimal. The controller 62 generates the drive signal bycontrolling (adjusting) the applied voltage independently of the appliedfrequency so that the supplied measured load angle approaches the targetload angle, and thereby the optimal efficiency can be achieved.

Further, in the present embodiment, frequency control is conducted in anouter loop. That is, a room temperature sensor 80 measures the roomtemperature, and supplies it to the host CPU 50. The host CPU 50controls (adjusts) the frequency command in accordance with the suppliedroom temperature.

Further, it is noted that a compressor, etc. may be employed as the load70.

FIG. 16 is a comparison chart of vector control with torque control(inverter) of the present embodiment.

FIG. 17 is a diagram showing one example of an optimal efficiency datatable obtained as a result of conducting a load test on a motor. We willexplain this further later.

The motor rotor equation of motion described here is defined usingstationary cylindrical coordinate system, which is employed throughoutthe present analysis. The stator winding located inside the motor ismagnetized by a digitalized sinusoidal wave from the driving inverter.Ferrite magnets (permanent magnets) are attached to the inside of themotor rotor forming the motor outer shell, and are magnetized bysinusoidal waveform as well.

In this analysis, we initially assume a two-pole/six slot motor andexpand it onto the 2-D plane. For simplicity, we assume a motorconfiguration having no salient poles. Further, the analysis isperformed for the U phase as the representative axis. FIG. 18 shows asynchronous motor model and its 2-D expansion view used in the presentembodiment.

In FIG. 18, assuming the moment when the armature current in the U phaseis maximum, the rotating magnetic flux ϕa generated by the armaturecurrent (armature reaction magnetic flux) which includes magnetic fluxesgenerated by the V phase and W phase currents is on the winding axis ofthe U phase. By noting that the rotating magnetic flux is rotating atthe synchronous speed toward the CCW (counterclockwise) direction in thedirection of the phase order, the electromotive force Ea=−jXal generatedby this magnetic flux is delayed from the current I by π/2. When this istreated as a voltage drop, the sign changes and it becomes advanced fromI by π/2.

The voltage drop jxlI by the armature leakage reactance is also advancedfrom I by π/2, and the sum of the two voltage drops j(Xa+xl) I=jXsl=thesynchronous reactance drop is also advanced from I by π/2.

If the center of the magnetic pole at the moment when the current of theU phase is maximum is at the position which is delayed from the windingaxis of the U phase by as shown in FIG. 18, the electromotive force E(We assume that the direction opposite to I is positive.) generated inthe U phase by the rotation of the magnetic flux ϕm by the magnetic poleat the synchronous speed toward the CCW direction in the direction ofthe phase order is delayed from jXsl in phase by ζ.

Further, when the armature winding and inverter lead resistance r isconsidered, the winding resistance equivalent magnetic flux ϕr by thisis in phase with I, and is delayed from ϕa by π/2.

The vector summation of E, rl and jXsl must be the power supply voltageV applied to the U phase, and therefore the vector diagram of the Uphase is obtained as shown in FIG. 19.

By referring to FIG. 20, we start the analysis using the followingequation of electromagnetic induction.V=dϕe(θ)/dt=pϕe(θ)  (1)

where

V=V1 sin (θ);

V1: Maximum voltage applied from the inverter at the U phase terminal;

θ: Rotational angle from the U phase terminal voltage axis Ue, CCWpositive;

ϕe=−ϕe1 cos (θ): Armature magnetic flux at a rotational angle θ

ϕe1: Positive maximum value of the armature magnetic flux induced by theU phase terminal voltage; and

p: Differentiation operator.

The equation (1) indicates that the phase of the armature magnetic fluxϕe is delayed from the phase of the terminal voltage V by π/2.

In other words, the armature magnetic flux ϕe at a phase angle of θ fromthe U phase terminal voltage axis Ue isϕe=−ϕe1 cos (θ)  (2)

In order to deal with the interaction with the permanent magnet magneticflux, it is convenient to define the armature magnetic flux axis Um byrotating the base axis of the armature magnetic flux ϕe in the clockwisedirection by π/2 in the x-y plane from the U phase terminal voltage axisUe. By using this redefinition, the armature magnetic flux ϕe can bewritten as.ϕe=ϕe1 sin (θ)  (3)

where

θ: Rotational angle from the U phase armature magnetic flux axis Um, CCWpositive.

A further manipulation of the equation (1) results inV=pϕe=jωϕe  (4)

Rewriting equation (4) for ϕe results inϕe=V/jω=−jV/ω  (5)

Similarly to the equation (1), the following equation of electromagneticinduction can be defined.E=dϕm(θ)/dt=pϕm(θ)  (6)

where

E=E1 sin(θ);

E1: Maximum voltage induced by the permanent magnet;

θ: Rotational angle from the U phase permanent magnet voltage axis q,CCW positive;

ϕm=−ϕm1 cos(θ): Permanent magnet magnetic flux at a rotational angle θ

ϕm1: Positive maximum value of the permanent magnet magnetic fluxinduced by the permanent magnet;

p: Differentiation operator;

The equation (6) indicates that the phase of ϕm is delayed from thephase of the permanent magnet voltage E by π/2.

When the permanent magnet magnetic flux axis d is defined by rotatingthe base axis of the permanent magnet magnetic flux ϕm from thepermanent magnet voltage axis q by π/2 in the clockwise direction, thepermanent magnet magnetic flux ϕm can be written asϕm=ϕm1 sin (θ)  (7)

where

θ: Rotational angle from the U phase permanent magnet magnetic flux axisd, CCW positive.

Similarly to the equation (5),ϕm=−jE/ω  (8)

When considering the winding resistance r of the armature windingincluding the inverter lead, the winding resistance equivalent magneticflux ϕr is defined as follows.ϕr=−jrl/ω  (9)

By referring to FIG. 19, we will analyze the following conventionalmotor circuit equation.V=E+rl+jXsl  (10)

where

r: Winding resistance;

Xs: Synchronous Reactance=ωLs;

Ls: Synchronous Inductance.

Multiplying both sides of equation (10) by −j/ω, it becomes

$\begin{matrix}{\begin{matrix}{{{- j}\;{V/\omega}} = {{{- {jE}}/\omega} - {{jr}\;{l/\omega}{⫬ {{+ {Xsl}}/\omega}}}}} \\{= {{{- {jE}}/\omega} - {{jrl}/\omega} + {Lsl}}}\end{matrix}\quad} & (11)\end{matrix}$

Substituting equations (5), (8) and (9) into equation (11) yieldsϕe=ϕm+ϕr+LsI  (12)

Newly defining the synchronous inductance magnetic flux ϕa as follows.ϕa=LsI  (13)

The equation (12) further turns to the equation (14) as a vectorrelationship of magnetic fluxes.ϕe=ϕm+ϕr+ϕa  (14)

The relationship of the equation (14) is illustrated as a vector diagramin FIG. 21.

Traditionally, we have the following motor torque equation,

$\begin{matrix}{\begin{matrix}{T = {k\;\left\{ {{V}\mspace{14mu}{{E}/\left( {\omega\;{Xs}} \right)}} \right\}\mspace{14mu}\sin\mspace{14mu}\delta}} \\{= {k\left\{ {V\; 1E\;{1/\left( {\omega\;{Xs}} \right)}} \right\}\mspace{14mu}\sin\mspace{14mu}\delta}}\end{matrix}{\quad\quad}} & (15)\end{matrix}$

where

k=3P/2: Constant number;

P: Number of motor poles;

V1: Maximum voltage applied from the inverter at the U phase terminal;

E1: Maximum voltage induced by the permanent magnet;

ω: Motor rotation angular speed.

Substituting the equation of the synchronous reactance Xs=ωLs which isdefined in equation (10) in equation (15) results in

$\begin{matrix}{\begin{matrix}{T = {k\left\{ {V\; 1E\;{1/\left( {{\omega 2}\;{Ls}} \right)}} \right\}\mspace{14mu}\sin\mspace{14mu}\delta}} \\{\left. \left. {= {k\left\{ {V\;{1/\omega}} \right)*{\left( {E\;{1/\omega}} \right)/{Ls}}}} \right) \right\}\mspace{14mu}\sin\mspace{14mu}\delta}\end{matrix}\quad} & (16)\end{matrix}$

where|ϕe|=ϕe1=V1/ω  (17)|ϕm|=ϕm1=E1/ω  (18)

Substituting equations (17) and (18) in equation (16) and arranging itresult inT=k|ϕe∥ϕm|sin δ/Ls  (19)

where

δ: Load angle=Included angle between the permanent magnet magnetic fluxaxis and the armature magnetic flux axis

Equation (19) indicates the motor torque is proportional to the areasurrounded by the oblique sides ϕe and ϕm and their included angle δ.This motor torque equation is illustrated in FIG. 22.

From this, it is noted that the motor torque is approximatelyproportional to the load angle δ when the load angle Δ is small.

The maximum value of the permanent magnet magnetic flux induced by thepermanent magnet is given and unchangeable. However, the presentinverter can precisely control the voltage value of the applied voltage(i.e. the terminal voltage of an armature winding) independently fromthe applied frequency. Therefore, it can realize the optimum efficiencyby changing the magnitude of the armature magnetic flux in accordancewith the speed and the magnitude of the load of the motor.

The load angle measurement and control of the present embodiment are notalways executed during the entire motor drive. They are operationalwithin certain motor speed ranges in which the motor has entered into asteady operation. For the motor drive in a transient state of increasingor decreasing the speed of the motor, an open loop control is executedby fully utilizing the character of a PCWM scheme employed by thepresent embodiment. The frequency of the load angle measurement andcontrol of the present embodiment may be control of an extremely longinterval such as on a “minute” basis. However, at the time of detectingthe load angle, real time processing of a short interval by a countingsignal PCK which is output from the PCWM signal encoder, describedlater, is preferable.

A 24-poles/18-slots external rotor type motor is preferable forpractical use. A method for measuring the motor load angle for suchmotor will be explained by using FIG. 23 which shows one embodiment.

The rotational angle of the rotor (electric angle) θ during one cycle ofthe driving sinusoidal wave of the motor isθ=2π/(24/2)=π/6  (20)

As the above-described equation (1) indicates that the armature magneticflux axis Um is delayed from the armature voltage axis Ue by ¼wavelength, when the equation (20) is multiplied by this value, theincluded angle between the two axes is given as follows:¼θ=π/(6*4)=(π/24=7.5°  (21)

In the present embodiment, the motor load angle is measured by definingthe armature magnetic flux axis Um at the position which is delayed fromthe armature voltage axis Ue by π/2 (¼ wavelength) in electric angle,and measuring the phase difference between the armature magnetic fluxand the permanent magnet magnetic flux relative to the armature magneticflux axis Um.

The measurement of the motor load angle is made by converting thesinusoidal wave of each of the armature magnetic flux and the permanentmagnet magnetic flux to 50% duty on/off signal, and measuring the phasedifference between the signals after the conversion. Thereby, an A/Dconverter for measuring an amplitude becomes unnecessary, and a signalprocessing circuit which is tolerant to external noises can be realized.Further, in contrast with the first carrier (CK3 described later) usedin the PCWM scheme which has a constant frequency, the period of thesine wave of the voltage applied to the motor changes in accordance withthe drive frequency, and therefore by using the second carrier (thecounting signal PCK described later) which is synchronized to the sinewave frequency, what percentage of the wavelength at the current drivefrequency the magnitude of the load angle becomes, is digitally counted.

In FIG. 23, the sinusoidal wave of the armature magnetic flux isconverted to an on/off signal δD representing the magnitude of thearmature magnetic flux. The conversion is conducted inside the PCWMsignal encoder 116, which is described later. That is, the sinusoidalwave is converted to a signal which becomes on when the magnitude of thearmature magnetic flux is zero or more, and off when the magnitude ofthe armature magnetic flux is less than zero. Here, the ON signalindicates the N pole. Further, the sinusoidal wave of the permanentmagnet magnetic flux is detected at the Hall sensor installed on theHall sensor PCB (Print Circuit Board), and it is converted to an on/offsignal δH representing the magnitude of the permanent magnet magneticflux. That is, the sinusoidal wave is converted to a signal whichbecomes on when the magnitude of the permanent magnet magnetic flux iszero or more, and off when the magnitude of the permanent magnetmagnetic flux is less than zero. Here, the ON signal indicates the Spole. In the present embodiment, both δD and δH are made to be 50% dutyon/off signals.

In FIG. 23, the sinusoidal wave table position means the table positionof FIG. 3. Thus, in one wavelength (one period) of the drive frequency,the PCWM signal encoder outputs 720 pulses as the counting signal. Forexample, as a result of digital measurement, if the phase difference(measured load angle δL) between the armature magnetic flux and thepermanent magnet magnetic flux is 36 of the above-described pulses, thisis 5% (=36/720) of the wavelength, and therefore the measured load angleδL is 18° (=360°×0.05).

When the number of rotations of the drive motor is given, there existsthe load angle value which gives optimum efficiency for the varying loadmagnitude. A model follower adaptive control method is employed byadjusting the motor applied voltage to have the value of the countedload angle become the ideal value stored in the table. For this purpose,a load test is conducted on the motor in advance, to get the table shownin FIG. 17. The relationship between the voltage command VC and thetarget load angle δT shown in FIG. 17 is prepared for each of thecertain frequency ranges, and stored in the target load angle table 104.Thus, the target load angle table 104 stores the target load angle (δT)to be targeted for the frequency (FC) and the voltage (VC) applied tothe permanent magnet motor 64.

If the present load angle measurement system is likened to measurementof the passing time of a train passing at a railroad crossing, itresembles conducting fixed point observation of the time difference fromthe closing of the crossing gate to the arrival of the train. That is,measurement of the delay time of the permanent magnet magnetic flux axisfrom the armature magnetic flux axis is understood as the phasedifference between the respective pulse trains of the armature magneticflux and the permanent magnet magnetic flux which have been converted to50% duty.

By referring to FIG. 24 which shows one embodiment of the load anglecontrol, further detailed explanation will be given. FIG. 24specifically shows the constitution of the permanent magnet motor drivesystem 60 of the present embodiment shown in FIG. 15.

In the present embodiment, the load angle sensor 66 and the load anglecontrol block 101 perform the load angle measurement and control.However, as described previously, the load angle measurement and controlare not always done during the entire motor drive. They are performedwithin certain motor speed ranges in which the motor enters into steadyoperation.

The load angle control block 101 generates the voltage command VC basedon the frequency command FC and the measured load angle δL to controlthe load angle.

The frequency command FC supplied from the outside of FIG. 24, and thevoltage command VC which is calculated on real time inside the figureand described later, are input to a target load angle determinationblock 102 inside the load angle control block 101. The target load angledetermination block 102 determines the target load angle δT based on thefrequency command FC and the voltage command VC by referring to thetarget load angle table 104. The target load angle table 104 is a set ofvalues of load angles at which the efficiency becomes optimal, and whichare obtained by conducting load test on the motor in advance. The targetload angle table 104 is given in the format shown in FIG. 17 asdescribed previously.

The load angle sensor 66 in FIG. 24 subtracts the Hall sensor phasesignal (signal representing the phase of the Hall sensor, i.e., phase ofthe permanent magnet magnetic flux) δH output from the Hall sensor 136,from the armature magnetic flux phase signal (signal representing thephase of the armature magnetic flux) δD output from the PCWM signalencoder 116, which is described later, to obtain the measured load angleδL as the output.

The load angle sensor 66 outputs the measured load angle δL of whichmeasurement is made by the method described in FIG. 23. The load anglesensor 66 may output the measured load angle δL after averaging it. Themeasured load angle δL output from the load angle sensor 66 is theninput to a load angle error calculator 106.

The load angle error calculator 106 subtracts the measured load angle δLfrom the target load angle δT to obtain a load angle error δE. The loadangle error δE is input to a voltage command accumulator 112 inside avoltage command generator 107.

The voltage command generator 107 generates a voltage command VC. Thevoltage command generator 107 comprises a base voltage determinationblock 108, a V/F base voltage table 110, and the voltage commandaccumulator 112.

The frequency command FC is input to the base voltage determinationblock 108. The base voltage determination block 108 determines a basevoltage VB by referring to the V/F base voltage table 110. The V/F basevoltage table 110 is a one obtained by conducting a load test on themotor in advance, and given in the format shown in FIG. 25. The basevoltage VB output from the base voltage determination block 108, isinput to the voltage command accumulator 112 having anaddition/subtraction/storage function.

At the time of entering into the load angle measurement and controlloop, the voltage command accumulator 112 outputs the base voltage VB asthe initial value of the voltage command VC, to the target load angledetermination block 102 and the PCWM encoder 116. Thereafter, thevoltage command accumulator 112 receives the load angle error δE fromthe load angle error calculator 106, and adjusts the voltage command VCbased on the load angle error δE. Specifically, when the load angleerror δE is plus, it means the measured load angle is less than thetarget load angle. Therefore, it works so that the voltage command VC isdecreased. Contrary, when the load angle error δE is minus, it means themeasured load angle is more than the target load angle. Therefore, itworks so that the voltage command VC is increased.

On the other hand, at the time of exiting from the load anglemeasurement and control loop, the voltage command accumulator 112continues to renew the value held by itself toward the value of the basevoltage VB so that the held value matches the base voltage VB in theend.

A PWM signal generator 114 comprises the PCWM signal encoder 116, a PCWMsignal decoder 128, and a sine wave 360° function table 120. Forexample, the PWM signal generator 114 can be realized as an LSI or anASIC. A logic part DC voltage 138 is supplied to the PWM signalgenerator 114. The PWM signal generator 114 generates a PWM signal basedon the frequency command FC and the voltage command VC.

Here, the PWM signal generator 114 can be configured similarly to theASIC 06 shown in FIGS. 1 and 2. However, in the present embodiment, theclocks CK1, CK3, CK4 and CK5 shown in FIG. 2 have been made eight timesgreater. That is, CK1, CK3, CK4 and CK5 have been made to be 1.6 μs,409.6 μs, 409.6 μs and 29.4912 ms, respectively. Further, until themachine frequency FM reaches the frequency command (command frequency)FC, the process shown in FIG. 6 is conducted, and load angle control isnot conducted. After the machine frequency FM reaches the frequencycommand (command frequency) FC, the voltage command VC is input as amachine voltage VM to the PCWM signal encoder 116, and load anglecontrol is conducted.

The PCWM signal encoder 116 receives the frequency command FC and thevoltage command VC as inputs, and receives data stored in the sine wave360° function table 120 shown in FIG. 3 as a write signal 122 by a readsignal 118 of a prescribed period. The PCWM signal encoder 116 outputsan encoded PCWM signal 126 by real time processing the write signal 122and the information of the frequency command FC and the voltage commandVC.

The format of the sine wave 360° function table 120 is the same as theunit sine function table in FIG. 3. The table consists of 720 8-bitbinary signals. 720 is selected as a multiple of 6. In the presentembodiment, a three phase permanent magnet type motor is used, andtherefore it is preferable to select a multiple of 6.

A figure showing how the information of the sine wave 360° functiontable 120 is processed in the PCWM signal encoder 116, is the fractionalsine function numeric nf representing the instantaneous amplitude valueand the pulse width numeric pw of FIG. 4. It shows that after theinverter starts at the origin of the circle, the scanning speedincreases with the operation point moving toward the outside as it isaccelerated, and when it reaches the maximum speed, the operation pointcirculates on the outer periphery of the circle.

A figure showing how the fractional sine function numeric nf in FIG. 4is converted to the pulse width numeric pw, is the relationship betweenthe pulse width numeric pw and the fractional sine function numeric nfof FIG. 5. However, as described above, in the present embodiment, CK1and CK3 are made as 1.6 μs and 409.6 μs, respectively.

As shown in the left side of FIG. 5, based on the calculation formulashown in FIG. 5, the encoded pulse width numeric pw in the unit PWMpulse interval is obtained, and this is output as the PCWM signal 126.

The PCWM signal decoder 128 decodes the PCWM signal 126 input from thePCWM signal encoder 116 on real time as a PWM signal 130, and outputs itto the inverter (gate drive) 132 of the next stage. The decoding methodof the PCWM signal decoder 128 is described previously with reference toFIGS. 10-14.

The inverter 132 generates a motor drive signal 134 based on the PWMsignal 130. The inverter 132 can be configured similarly to the gatedrive and power transistor circuitry 08 shown in FIGS. 1 and 2. Themotor drive signal 134 output from the inverter 132 drives the threephase permanent magnet motor 64 which is directly coupled to the load70. A main circuit DC voltage 140 is supplied to the inverter 132.

The controller 62 applies the voltage to the permanent magnet motor 64,and therefore it knows the state of the armature magnetic flux at eachtime. The PCWM signal encoder 116 of the controller 62 outputs, amongits outputs, the armature magnetic flux phase signal δD which is anon/off signal of 50% duty representing the magnitude (zero or more, orless than zero) of the sine wave signal of the armature magnetic flux.

In the case of FIG. 24, the Hall sensor 136 is installed on the statorside of the three phase permanent magnet motor 64 (see also FIG. 23).The Hall sensor 136 detects the sine wave signal generated by therotation of the three phase permanent magnet motor 64, converts thesignal to the Hall sensor phase signal δH which is an on/off signal of50% duty representing the magnitude (zero or more, or less than zero) ofthe signal by a comparator in the Hall sensor 136, and outputs it.

The PCWM signal encoder 116 outputs the counting signal (read signal)PCK which becomes on/off at every occurrence of the write signal 122.The counting signal PCK outputs as many pulses as the number of datastored in the sine wave 360° function table 120 during one period of thedrive frequency signal regardless of the magnitude of the drivefrequency. This can be called the second carrier which is synchronizedto the period of the drive frequency. When the phase difference betweenthe armature magnetic flux phase signal δD and the Hall sensor phasesignal δH is digitally measured, the phase difference becomes aneffective means as an index showing the ratio to the wavelength of thedrive frequency signal.

The load angle sensor 66 receives the armature magnetic flux phasesignal δD and the Hall sensor phase signal δH as inputs, counts thephase difference between the two by the counting signal PCK, and outputsthe resulting number of counts as the measured load angle δL.

In this way, it is possible to count digitally what percentage of thewavelength at the current drive frequency the magnitude of the loadangle becomes. Namely, the inverter 132 of the controller 62 applies thesine wave voltage to the permanent magnet motor 64. Further, the phaseof the voltage is expressed in n=720 ways (see FIGS. 3 and 4). Further,the PCWM signal encoder 116 of the controller 62 sends n=720 countingsignals PCK (pulses) to the load angle sensor 66 during one period (onewavelength) of the voltage. Then, the load angle sensor 66 measures theload angle by measuring the number of pulses which corresponds to thephase difference between the armature magnetic flux and the permanentmagnet magnetic flux corresponds to. Thus, the load angle sensor 66 cancount digitally what percentage of the wavelength at the current drivefrequency the magnitude of the load angle becomes.

In the present embodiment, the phase of the voltage is expressed inn=720 ways. However, another value (integer equal to or greater than 2)can be employed as n. Here, as the value of n, 6 or more is preferable.Specifically, a multiple of 6 which is 6 or more is preferable.

Further, in the present embodiment, a Hall sensor is used as a permanentmagnet magnetic flux sensor detecting the permanent magnet magneticflux, but another permanent magnet magnetic flux sensor may be used.

Second Embodiment

In the above-described first embodiment, control is conducted based onload angle, but in the second embodiment of the present disclosure,control is conducted based on power factor angle.

FIG. 26 illustrates a permanent magnet motor drive system (synchronousmotor drive system) in the second embodiment of the present disclosure,and represents a circuit block diagram for an efficiency control ofeither an air conditioner or a fan. The permanent magnet motor drivesystem 60 of the present embodiment comprises a controller 62, apermanent magnet motor 64, and a power factor angle sensor 67. In thepresent embodiment, as a load 70, for example, an air conditioner or afan is assumed. When a user inputs a room temperature command to a hostCPU 50, the host CPU 50 inputs a frequency command which depends on theroom temperature command, to the controller 62 inside the permanentmagnet motor drive system 60. The controller 62 generates a drive signalwhich depends on the frequency command, and supplies it to the permanentmagnet motor 64. The permanent magnet motor 64 operates in accordancewith the drive signal, and supplies a speed and a torque to the load 70.The load (air conditioner or fan) 70 operates in accordance with thespeed and the torque, and the room temperature changes.

In the present embodiment, a power factor angle is controlled byregulating a voltage in an inner loop. That is, the power factor anglesensor 67 measures the power factor angle of the permanent magnet motor64, and supplies it to the controller 62. The controller 62 generatesthe drive signal based on the frequency command and the measured powerfactor angle. Here, for the number of rotations of the permanent magnetmotor 64, in accordance with the magnitude of the load, there exists thevalue of the power factor angle (target power factor angle) at which theefficiency of the permanent magnet motor 64 becomes optimal. Thecontroller 62 generates the drive signal by controlling (adjusting) theapplied voltage independently of the applied frequency so that thesupplied measured power factor angle approaches the target power factorangle, and thereby the optimal efficiency can be achieved.

Further, in the present embodiment, frequency control is conducted in anouter loop. That is, a room temperature sensor 80 measures the roomtemperature, and supplies it to the host CPU 50. The host CPU 50controls (adjusts) the frequency command in accordance with the suppliedroom temperature.

Further, it is noted that a compressor, etc. may be employed as the load70.

FIG. 27 is a diagram showing one example of an optimal efficiency datatable obtained as a result of conducting a load test on a motor. We willexplain this further later.

The power factor angle measurement and control of the present embodimentare not always executed during the entire motor drive. They areoperational within a certain motor speed range in which the motor hasentered into steady operation. For the motor drive in a transient stateof increasing or decreasing the speed of the motor, an open loop controlis executed by fully utilizing the character of a PCWM scheme employedby the present embodiment. The frequency of the power factor anglemeasurement and control of the present embodiment may be control of anextremely long interval such as on a “minute” basis. However, at thetime of detecting the power factor angle, real time processing of ashort interval by a counting signal PCK output from the PCWM signalencoder, described later, is preferable.

A 24-poles/18-slots external rotor type motor is preferable forpractical use. A method for measuring the motor power factor angle forsuch motor will be explained by using FIGS. 28 and 29 which show oneembodiment.

The measurement of the motor power factor angle is conducted bymeasuring the phase difference between a terminal voltage (In thepresent embodiment, the U phase terminal voltage is used as arepresentative of the three phases.) of the motor and a terminal current(In the present embodiment, the U phase terminal current is used as arepresentative of the three phases.) of the motor. Terminals of themotor and terminals of an inverter are connected each other. Therefore,the terminal voltage and the terminal current of the motor are identicalto the terminal voltage and the terminal current of the inverter,respectively. In the present embodiment, a current sensor 144 detectsthe terminal current of the inverter, and thereby the terminal currentof the motor is detected. Specifically, the measurement of the motorpower factor angle is made by converting the sine wave of each of theterminal voltage and the terminal current to 50% duty on/off signal, andmeasuring the phase difference between the signals after the conversion.Thereby, an A/D converter for measuring an amplitude becomesunnecessary, and a signal processing circuit which is tolerant toexternal noises can be realized. Further, in contrast with the firstcarrier (CK3 described later) used in the PCWM scheme which has aconstant frequency, the period of the sine wave of the voltage appliedto the motor changes in accordance with a drive frequency, and thereforeby using the second carrier (the counting signal PCK described later)which is synchronized to the sine wave frequency, what percentage of thewavelength at the current drive frequency the magnitude of the powerfactor angle becomes, is digitally counted.

In FIGS. 28 and 29, the sine wave of the terminal voltage is convertedto an on/off signal δV representing the magnitude of the terminalvoltage. The conversion is conducted inside a PCWM signal encoder 116,which is described later. That is, the sine wave is converted to asignal which becomes on when the magnitude of the terminal voltage iszero or more, and off when the magnitude of the terminal voltage is lessthan zero. Secondly, the sine wave of the terminal current is convertedto an on/off signal δI representing the magnitude of the terminalcurrent. The conversion is conducted inside the current sensor 144,which is described later. That is, the sine wave is converted to asignal which becomes on when the magnitude of the terminal current iszero or more, and off when the magnitude of the terminal current is lessthan zero. In the present embodiment, both δV and δI are made to be 50%duty on/off signals.

FIG. 28 shows a case where the voltage phase is advanced from thecurrent phase. In FIG. 28, the sinusoidal wave table position isidentical to the table position of FIG. 3. Thus, in one wavelength (oneperiod) of the drive frequency, the PCWM signal encoder outputs 720pulses as the counting signal. For example, as a result of digitalmeasurement, if the phase difference (measured power factor angle δP)between the voltage phase and the current phase is 36 of theabove-described pulses, this is 5% (=36/720) of the wavelength, andtherefore the measured power factor angle δP is 18° (=360°×0.05).

FIG. 29 shows a case where the voltage phase is delayed from the currentphase. In FIG. 29, the sinusoidal wave table position is identical tothe table position of FIG. 3. Thus, in one wavelength (one period) ofthe drive frequency, the PCWM signal encoder outputs 720 pulses as thecounting signal. For example, as a result of digital measurement, if thephase difference (measured power factor angle δP) between the voltagephase and the current phase is 36 of the above-described pulses, this is5% (=36/720) of the wavelength, and therefore the measured power factorangle δP is −18° (=−360°×0.05).

There exists the power factor angle value at which the efficiencybecomes optimal for the varying load magnitude at the given number ofrotations of the drive motor. A model follower adaptive control methodis employed by adjusting the motor applied voltage to have the value ofthe counted power factor angle become the ideal value stored in thetable. For this purpose, a load test is conducted on the motor inadvance, to get the table shown in FIG. 27. The relationship between thevoltage command VC and the target power factor angle δS shown in FIG. 27is prepared for each of the certain frequency ranges, and stored in thetarget power factor angle table 105. Thus, the target power factor angletable 105 stores the target power factor angle (δS) to be targeted forthe frequency (FC) and the voltage (VC) applied to the permanent magnetmotor 64.

As shown in FIG. 23, in use for a fan or a blower, a Hall sensor can beinstalled on the motor stator. However, for an air conditionercompressor application, the Hall sensor cannot be installed due toenvironmental restriction inside the compressor. Therefore, the currentsensor shown in FIG. 30 is employed instead.

Next, by referring to FIG. 30 which shows one embodiment of power factorangle control, further detailed explanation will be given. FIG. 30specifically shows the constitution of the permanent magnet motor drivesystem 60 of the present embodiment shown in FIG. 26.

In the present embodiment, the power factor angle sensor 67 and thepower factor angle control block 111 perform the power factor anglemeasurement and control. However, as described previously, the powerfactor angle measurement and control are not always done during theentire motor drive. They are performed within certain motor speed rangesin which the motor enters into steady operation.

The power factor angle control block 111 generates the voltage commandVC based on the frequency command FC and the measured power factor angleδP to control the power factor angle.

The frequency command FC supplied from the outside of FIG. 30, and thevoltage command VC which is calculated on real time inside the figureand described later, are input to a target power factor angledetermination block 103 inside the power factor angle control block 111.The target power factor angle determination block 103 determines thetarget power factor angle δS based on the frequency command FC and thevoltage command VC by referring to the target power factor angle table105. The target power factor angle table 105 is a set of values of powerfactor angles at which the efficiency becomes optimal, and which areobtained by conducting load test on the motor in advance. The targetpower factor angle table 105 is given in the format shown in FIG. 27 asdescribed previously.

The power factor angle sensor 67 in FIG. 30 subtracts the current phasesignal δI output from the current sensor 144, from the voltage phasesignal δV output from the PCWM signal encoder 116, which is describedlater, to obtain the measured power factor angle δP as the output.

The power factor angle sensor 67 outputs the measured power factor angleδP of which measurement is made by the method described in FIGS. 28 and29. That is, it subtracts the current phase signal (signal representingthe phase of the terminal current) δI, from the voltage phase signal(signal representing the phase of the terminal voltage) δV, to obtainthe output which is the measured power factor angle δP. The measuredpower factor angle δP is output to a power factor angle error calculator109.

The power factor angle error calculator 109 subtracts the measured powerfactor angle δP from the target power factor angle δS to obtain a powerfactor angle error δF. The power factor angle error δF is input to avoltage command accumulator 112 inside a voltage command generator 107.

The voltage command generator 107 generates a voltage command VC. Thevoltage command generator 107 comprises a base voltage determinationblock 108, a V/F base voltage table 110, and the voltage commandaccumulator 112.

The frequency command FC is input to the base voltage determinationblock 108. The base voltage determination block 108 determines a basevoltage VB by referring to the V/F base voltage table 110. The V/F basevoltage table 110 is a one obtained by conducting a load test on themotor in advance, and given in the format shown in FIG. 25. The basevoltage VB output from the base voltage determination block 108, isinput to the voltage command accumulator 112 having anaddition/subtraction/storage function.

At the time of entering into the power factor angle measurement andcontrol loop, the voltage command accumulator 112 outputs the basevoltage VB as the initial value of the voltage command VC, to the targetpower factor angle determination block 103 and the PCWM encoder 116.Thereafter, the voltage command accumulator 112 receives the powerfactor angle error δF from the power factor angle error calculator 109,and adjusts the voltage command VC based on the power factor angle errorδF. Specifically, when the power factor angle error δF is plus, it meansthe voltage phase is delayed from the current phase beyond the targetvalue, and the load is light. Therefore, it works so that the voltagecommand VC is decreased. Contrary, when the power factor angle error δFis minus, it means the voltage phase is advanced from the current phasebeyond the target value, and the load is heavy. Therefore, it works sothat the voltage command VC is increased.

On the other hand, at the time of exiting from the power factor anglemeasurement and control loop, the voltage command accumulator 112continues to renew the value held by itself toward the value of the basevoltage VB so that the held value matches the base voltage VB in theend.

A PWM signal generator 114 comprises the PCWM signal encoder 116, a PCWMsignal decoder 128, and the sine wave 360° function table 120. Forexample, the PWM signal generator 114 can be realized as an LSI or anASIC. A logic part DC voltage 138 is supplied to the PWM signalgenerator 114. The PWM signal generator 114 generates a PWM signal basedon the frequency command FC and the voltage command VC.

Here, the PWM signal generator 114 can be configured similarly to theASIC 06 shown in FIGS. 1 and 2. However, in the present embodiment, theclocks CK1, CK3, CK4 and CK5 shown in FIG. 2 are made eight timeslonger. Namely, CK1, CK3, CK4 and CK5 are made as 1.6 μs, 409.6 μs,409.6 μs and 29.4912 ms, respectively. Further, until the machinefrequency FM reaches the frequency command (command frequency) FC, theprocess shown in FIG. 6 is conducted, and the power factor angle controlis not conducted. After the machine frequency FM reaches the frequencycommand (command frequency) FC, the voltage command VC is input as themachine voltage VM to the PCWM signal encoder 116, and power factorangle control is conducted.

The PCWM signal encoder 116 receives the frequency command FC and thevoltage command VC as inputs, and receives data stored in the sine wave360° function table 120 shown in FIG. 3 as a write signal 122 by a readsignal 118 of a prescribed period. The PCWM signal encoder 116 outputsan encoded PCWM signal 126 by real time processing the write signal 122and the information of the frequency command FC and the voltage commandVC.

The format of the sine wave 360° function table 120 is the same as theunit sine function table in FIG. 3. The table consists of 720 8-bitbinary signals. 720 is selected as a multiple of 6. In the presentembodiment, a three phase permanent magnet type motor is used, andtherefore it is preferable to select a multiple of 6.

A figure showing how the information of the sine wave 360° functiontable 120 is processed in the PCWM signal encoder 116, is the fractionalsine function numeric nf representing the instantaneous amplitude valueand the pulse width numeric pw of FIG. 4. It shows that after theinverter starts at the origin of the circle, the scanning speedincreases with the operation point moving toward the outside as it isaccelerated, and when it reaches the maximum speed, the operation pointcirculates on the outer periphery of the circle.

A figure showing how the fractional sine function numeric nf in FIG. 4is converted to the pulse width numeric pw, is the relationship betweenthe pulse width numeric pw and the fractional sine function numeric nfof FIG. 5. However, as described above, in the present embodiment, CK1and CK3 are made as 1.6 μs and 409.6 μs, respectively.

As shown in the left side of FIG. 5, based on the calculation formulashown in FIG. 5, the encoded pulse width numeric pw in the unit PWMpulse interval is obtained, and this is output as the PCWM signal 126.

The PCWM signal decoder 128 decodes the PCWM signal 126 input from thePCWM signal encoder 116 on real time as a PWM signal 130, and outputs itto the next stage inverter (gate drive) 132. The decoding method of thePCWM signal decoder 128 is described previously with reference to FIGS.10-14.

The inverter 132 generates a motor drive signal 134 based on the PWMsignal 130. The inverter 132 can be configured similarly to the gatedrive and power transistor circuitry 08 shown in FIGS. 1 and 2. Themotor drive signal 134 output from the inverter 132 drives the threephase permanent magnet motor 64 which is directly coupled to the load70. A main circuit DC voltage 140 is supplied to the inverter 132.

The controller 62 applies the voltage to the permanent magnet motor 64,and therefore it knows the state of the terminal voltage at each time.The PCWM signal encoder 116 of the controller 62 outputs, among itsoutputs, the voltage phase signal δV which is an on/off signal of 50%duty representing the sine wave signal magnitude (zero or more, or lessthan zero) of the terminal voltage.

In the case of FIG. 30, the current sensor 144 is installed on theinverter board of the controller 62, and is coupled to the drive signal134 of the three phase permanent magnet motor 64 without contact by theHall effect. The current sensor 144 converts the drive current (terminalcurrent) signal of the three phase permanent magnet motor 64 to thecurrent phase signal δI which is an on/off signal of 50% dutyrepresenting the magnitude (zero or more, or less than zero) of thesignal by a comparator in the current sensor 144, and outputs thecurrent phase signal δI.

The PCWM signal encoder 116 outputs the counting signal (read signal)PCK which becomes on/off at every occurrence of the write signal 122.The counting signal PCK outputs as many pulses as the number of datastored in the sine wave 360° function table 120 during one period of thedrive frequency signal regardless of the magnitude of the drivefrequency. This can be called the second carrier which is synchronizedto the period of the drive frequency. When the phase difference betweenthe voltage phase signal δV and the current phase signal δI is digitallymeasured, the phase difference becomes an effective means as an indexshowing the ratio to the wavelength of the drive frequency signal.

The power factor angle sensor 67 receives the voltage phase signal δVand the current phase signal δI as inputs, counts the phase differencebetween the two by the counting signal PCK, and outputs the resultingnumber of counts as the measured power factor angle δP.

In this way, it is possible to count digitally what percentage of thewavelength at the current drive frequency the magnitude of the powerfactor angle becomes. Namely, the inverter 132 of the controller 62applies the sine wave voltage to the permanent magnet motor 64. Further,the phase of the voltage is expressed in n=720 ways (see FIGS. 3 and 4).Further, the PCWM signal encoder 116 of the controller 62 sends n=720counting signals PCK (pulses) to the power factor angle sensor 67 duringone period (one wavelength) of the voltage. Then, the power factor anglesensor 67 measures the power factor angle by measuring the number ofpulses which corresponds to the phase difference between the terminalvoltage and the terminal current corresponds to. Thus, the power factorangle sensor 67 can count digitally what percentage of the wavelength atthe current drive frequency the magnitude of the power factor anglebecomes.

In the present embodiment, the phase of the voltage is expressed inn=720 ways. However, another value (integer equal to or greater than 2)can be employed as n. Here, as the value of n, 6 or more is preferable.Specifically, a multiple of 6 which is 6 or more is preferable.

Others

In the above-described embodiments, a permanent magnet motor (threephase permanent magnet motor) is used as a synchronous motor, but thepresent disclosure can be applied to other synchronous motors. Further,an outer rotor type motor is used as a motor, but the present disclosurecan be applied to an inner rotor type motor.

Those having skill in this art will understand that many changes may bemade to the details of the above-described embodiments without departingfrom the underlying principles of the present disclosure. Therefore, thescope of the present invention is determined only by the claims.

The invention claimed is:
 1. A synchronous motor drive systemcomprising: a synchronous motor; a power factor angle sensor formeasuring a power factor angle of the synchronous motor; and acontroller for generating a drive signal based on an input frequencycommand and the measured power factor angle, and supplying the drivesignal to the synchronous motor, wherein the controller comprises apower factor angle control block for generating a voltage command basedon the frequency command and the measured power factor angle to controlthe power factor angle, and wherein the power factor angle control blockcomprises: a voltage command generator for generating the voltagecommand; a target power factor angle table storing a target power factorangle to be targeted for a frequency and a voltage applied to thesynchronous motor; a target power factor angle determination block fordetermining the target power factor angle based on the frequency commandand the voltage command by referring to the target power factor angletable; and a power factor angle error calculator for calculating a powerfactor angle error between the target power factor angle and themeasured power factor angle, wherein the voltage command generatoradjusts the voltage command to be generated based on the power factorangle error.
 2. The synchronous motor drive system as claimed in claim1, wherein the synchronous motor drive system further comprises aterminal current detection sensor for detecting a terminal current ofthe synchronous motor, and the power factor angle sensor measures aphase difference between a terminal voltage of the synchronous motor andthe terminal current to measure the power factor angle.
 3. Thesynchronous motor drive system as claimed in claim 2, wherein thecontroller sends a voltage phase signal representing a phase of theterminal voltage to the power factor angle sensor, the terminal currentdetection sensor sends a current phase signal representing a phase ofthe terminal current to the power factor angle sensor, and the powerfactor angle sensor measures the phase difference between the terminalvoltage and the terminal current based on the voltage phase signal andthe current phase signal.
 4. The synchronous motor drive system asclaimed in claim 3, wherein the controller sends, as the voltage phasesignal, an on/off signal representing a magnitude of the terminalvoltage, and the terminal current detection sensor sends, as the currentphase signal, an on/off signal representing a magnitude of the terminalcurrent.
 5. The synchronous motor drive system as claimed in claim 2,wherein the controller applies a sine wave voltage to the synchronousmotor, expresses a phase of the voltage in n ways (n is an integer equalto or greater than 2), and sends n pulses to the power factor anglesensor during one period of the voltage, and the power factor anglesensor measures the power factor angle by measuring the number of pulseswhich corresponds to the phase difference between the terminal voltageand the terminal current.
 6. The synchronous motor drive system asclaimed in claim 1, wherein the controller further comprises: a PWMsignal generator for generating a PWM signal based on the frequencycommand and the voltage command; and an inverter for generating thedrive signal based on the PWM signal.